Direct conversion polar transmitter for an rfid reader

ABSTRACT

A polar transmitter for an RFID reader and a system using the polar transmitter are disclosed. An RFID system according to at least some embodiments of the invention includes a polar transmitter, a receiver to receive responses from RFID tags, and a coupler connected to the polar transmitter, the receiver and one or more antennas. In at least some embodiments, the polar transmitter of the RFID system includes an envelope amplifier and a power amplifier. In some examples, a polar transmitter includes direct conversion of baseband data to provide angle modulation plus drive modulation. In addition to the envelope amplifier and power amplifier, the polar transmitter in such an example includes a quadrature modulator connected to the power amplifier to provide modulation for the transmitter output signal using a Cartesian input signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of and claims priority fromU.S. Pat. No. 10,540,526, entitled “Polar Transmittal Using Multi-PhaseBuck Converter,” which is a continuation-in-part of and claims priorityfrom U.S. Pat. No. 10,387,691, entitled “Polar Transmitter For an RFIDReader,” which is from a National Stage Entry of InternationalApplication No. PCT/US2014/068488, filed Dec. 4, 2014 and entitled“Polar Transmitter For an RFID Reader,” which claims priority to U.S.Provisional Application No. 61/937,789, filed Feb. 10, 2014 and entitled“Polar Transmitter For an RFID Reader,” each of which is herebyincorporated by reference herein.

BACKGROUND

Radio frequency identification (RFID) is used in a wide variety oflogistics, supply chain, manufacturing, and other applications. For someRFID protocols the reader may modulate commands onto a radio frequency(RF) carrier signal and the RFID tags respond to the modulated commands.The reader's modulation must include amplitude modulation because someRFID tag's commonly use envelope detection to decode the commands. Theamplitude modulation requires that the reader's RF power amplifier (RFPAor “power amplifier”) be linear enough to pass regulatory and protocolconformance testing. However, there is typically a trade-off betweenRFPA linearity and RFPA power efficiency. To reduce cost and packagesize, the high performance RFID reader design must jointly optimize RFPAlinearity and power efficiency. Many conventional RFID readers useclass-AB RFPA designs for which linearity and power efficiency may beconflicting properties.

SUMMARY

Embodiments of the present invention provide apparatus and methods foran improved RFID reader design using a polar transmitter. The apparatusaccording to embodiments of the invention can significantly improve thepower efficiency and reduce the cost of the RFID reader. Embodiments ofthe invention, for example, may achieve power efficiencies of 66% orbetter. This can yield significant reductions in dissipated power withcorresponding reductions in package size and cost as well as improveddevice reliability, since high temperatures due to dissipating power isoften a cause of electronics failure. For example, a reader requiring RFpower amplifier (RFPA) output of two watts using in a conventionalquasi-linear class AB design at 33% efficiency will dissipate four wattsof power internally. However, when using the polar transmitter designdisclosed herein with 66% efficiency the RFPA would only dissipate onewatt of power internally. The polar transmitter architecture disclosedherein can also eliminate expensive components such as the RF mixer ormodulator as used in a direct conversion transmitter.

An RFID system according to at least some embodiments of the inventionincludes a polar transmitter employing a switch mode power amplifier,most often class E, inverse class E, class F, or inverse class F. By aswitch mode power amplifier, what is meant is a circuit designed toconvert DC power or low frequency power as from an envelope amplifierinto RF power by approximating an ideal switch operation, e.g., byswitching on and off at the radio frequency. The on/off switching occursat the desired radio frequency thereby converting low frequency power toRF power.

In at least some embodiments of the invention the RF source whichcontrols the switch mode power amplifier is phase modulated using aphase modulator provided with a phase signal to generate a phasemodulated output,

u(t)=e·cos(ωt+p(t)).

In at least some embodiments of the invention the switch mode poweramplifier's power source is amplitude modulated using an envelopeamplifier to produce an amplitude modulated output,

u(t)=e(t)·cos(ωt).

In at least some embodiments of the invention both amplitude and phasemodulation are used to produce a modulated output

u(t)=e(t)·cos(ωt+p(t))=Re{e(t)·exp(jp(t))},

thus the name “polar transmitter”, since the output signal is generatedfrom a polar representation of the signal.

To achieve good spectral occupancy performance of the transmitter, themodulation employed in some embodiments may use OPR-ASK (offset phasereversal amplitude shift keying), which is PR-ASK with orthogonal offsetcarrier injection as disclosed in U.S. Pat. No. 9,813,115 B2 entitled,“RF System Using PR-ASK with Orthogonal Offset,” which is incorporatedherein by reference. The OPR-ASK modulation technique enables thespectrally efficient implementation of the polar modulation transmitterdescribed herein. In at least some embodiments, the system also includesa receiver to receive responses from RFID tags and a coupler connectedto the polar transmitter and the receiver to serve as the means to passthe transmitter output signal to one or more antennas and to pass theresponses to the receiver.

In at least some embodiments, the polar transmitter of the RFID systemincludes an envelope amplifier connected to the switch mode poweramplifier to provide an envelope signal and a phase modulator connectedto the switch mode power amplifier to phase modulate the switch modepower amplifier using a phase signal. The envelope amplifier inputsignal and the phase signal can be sampled data signals produced bymeans of a processor such as a digital signal processor. In such cases,the signals may be referred to as a sampled data envelope amplifiersignal and a sampled data phase signal. In some embodiments the envelopeamplifier can include or be powered by a switch mode power supply. Insome embodiments, the power supply and hence the envelope amplifier caninclude a multi-phase buck converter, though other types of switch modetopologies can alternatively be used.

In some examples, a polar transmitter includes direct conversion ofbaseband data to provide angle modulation plus drive modulation. Inaddition to an envelope amplifier and power amplifier, the polartransmitter in such an example includes a quadrature modulator connectedto the power amplifier to provide modulation for the transmitter outputsignal using a cartesian input signal. The polar transmitter in such anexample can include a driver amplifier connected to the power amplifierand can also include a DC blocking capacitor connected between thedriver amplifier and the power amplifier. An unmodulated RF source suchas a frequency synthesizer can be used to supply a continuous wavesignal to the quadrature modulator.

The polar transmitter according to at least some embodiments of theinvention operates in part by phase modulating the switch mode poweramplifier using the phase signal to produce the transmitter outputsignal based on the signal from the envelope amplifier. In someembodiments, the phase control is exercised by shifting the phase of thesignal, for example, using a phase shifter connected to a localoscillator. In some embodiments a direct digital synthesizer is used toimplement digital phase modulation. The envelope amplifier input signalcan be or can include a pulse width modulated (PWM) signal. In someembodiments a linear regulator can be used in the envelope amplifieralong with the switch mode power supply. The linear regulator and theswitch mode power supply can be connected either in series or inparallel.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram of an example operating environmentfor an RFID system with an amplitude modulated reader-to-tagcommunications link.

FIG. 2 is a block diagram of an RFID reader using a direct conversiontransmitter and high RF gain receiver.

FIG. 3 is a block diagram of an RFID reader using a polar modulationtransmitter and a low RF gain receiver for reduced cost.

FIG. 4 is a frequency domain plot showing the power spectrum for PR-ASK,OPR-ASK, and the AC components of the envelopes for those signals.

FIG. 5 is a block diagram of an RFID reader using a digitally controlledpolar modulation transmitter.

FIG. 6 is a block diagram of an RFID reader using a polar modulationtransmitter which employs a PWM envelope amplifier with linear regulatorassistance.

FIG. 7 is a detailed block diagram of an RFID polar transmitter with PWMand parallel linear regulator.

FIG. 8 is a diagram illustrating the communications link budget for apassive ultra-high frequency (UHF) backscatter RFID system.

FIG. 9 is a block diagram of a generic receiver lineup with multipleprocessing components in series, each having an associated gain (loss)and noise figure.

FIG. 10 is a high level block diagram of an RFID system using a polartransmitter wherein the transmitter and receiver functions may bephysically separated.

FIG. 11 is a block diagram of one embodiment of a polar transmitterusing a two-phase buck envelope amplifier.

FIG. 12 is a block diagram of one embodiment of a polar transmitterusing a direct conversion to provide angle modulation plus drivemodulation.

DETAILED DESCRIPTION

Embodiments of the present invention now will be described more fullyhereinafter with reference to the accompanying drawings, in whichembodiments of the invention are shown. This invention may, however, beembodied in many different forms and should not be construed as limitedto the embodiments set forth herein. Like numbers refer to like elementsthroughout.

Terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the invention. Asused herein, the singular forms “a”, “an” and “the” are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. It will be further understood that the terms “comprises” or“comprising,” when used in this specification, specify the presence ofstated features, steps, operations, elements, or components, but do notpreclude the presence or addition of one or more other features, steps,operations, elements, components, or groups thereof. Additionally,comparative, quantitative terms such as “above”, “below”, “less”,“more”, are intended to encompass the concept of equality, thus, “less”can mean not only “less” in the strictest mathematical sense, but also,“less than or equal to.”

Unless otherwise defined, all terms (including technical and scientificterms) used herein have the same meaning as commonly understood by oneof ordinary skill in the art to which this invention belongs. It will befurther understood that terms used herein should be interpreted ashaving a meaning that is consistent with their meaning in the context ofthis specification and the relevant art and will not be interpreted inan idealized or overly formal sense unless expressly so defined herein.It will also be understood that when an element is referred to as being“connected” or “coupled” to another element, it can be directlyconnected or coupled to the other element or intervening elements may bepresent. Similar or like components in different portions of an examplecircuit or flowchart can be referred using the adjectives “first,”“second,” etc.

This disclosure has to do with generating signals for RFID transmission.The conventional approach to generating modulated transmission signalsis to use digital signal processing (DSP) techniques which produce a“digital signal”, which is a quantized and sampled signal. These digitalsignals may be denoted such as x(n), where the argument is a variablesuch as n, m, or k. These variables represent the sample index,typically under a uniform sampling period T_(S). The digital signal ispassed from the DSP to a digital-to-analog converter (DAC) whichproduces a continuous time version of the signal, commonly using azero-order-hold representation at the DAC output. For band limitedsignal reproduction the DAC output is followed by a low pass or bandpass reconstruction filter which produces a continuous time, continuousamplitude version x(t) of the digital signal x(n), where we havereplaced the sampled time argument “n” with the continuous time argument“t”. The signal name “x” remains the same to indicate the sampled timeand continuous time signals represent the same signal, even though therewill generally be scaling differences and small time delays in the twosignal representations. This Nyquist sampling theory and practice ofsampled data systems is well known to those skilled in the art.

A block diagram of the operating environment of an example RFID reader1000 using amplitude modulated reader-to-tag link communications signal3000 is shown in FIG. 1. The reader 1000 is connected to one or moreantennas 2000, which radiate the modulated transmit signal 3000 to oneor more tags 4000. Some types of RFID tags decode commands from thereader which are encoded in the amplitude modulation of the reader's RFcarrier signal. The modulated reader-to-tag signal 3000 may also containphase modulation, but many types of tags only use the envelope fordecoding. In some cases the envelope modulation depth of the reader's RFcarrier is specified to be from 80% to 100%. Examples of this are theISO 18000-63 protocol and EPCGlobal C1G2 protocol, also informally knownas “Gen2”. The reader modulation typically uses either Manchester linecodes or Pulse Interval Encoding (PIE) line codes. These two line codesare both present in the ISO 18000-6 standards for RFID in the ultra-highfrequency (UHF) band. Note that although the example embodimentsdescribed herein are focused on UHF band RFID, the methods and apparatusdescribed in the following can readily be applied to the 2.45 GHzmicrowave band, or other RFID readers and protocols in the UHF ormicrowave bands.

In backscatter RFID systems the data communications is typically halfduplex. When finished modulating a command the reader 1000 willtypically leave its RF carrier signal active so that one or more tags4000 may backscatter modulate their response to the reader. If the tags4000 are passive then they use the RF signal from the reader 1000 forpower, while semi-passive tags have batteries to operate their circuitryinstead of harvesting power from the reader RF transmission. All thetags 4000, whether passive or semi-passive, use the RF carrier signalfrom the reader 1000 as the medium to communicate back to the reader viabackscatter modulation. The tags modulate their radar cross section tovary the amount of the reader's RF carrier which is reflected to thereader.

FIG. 2 illustrates an architecture for RFID reader 1000. The transmitter1300 uses direct conversion to generate the amplitude modulated RFsignal. In this design the in-phase 1210 and quadrature-phase 1220baseband signals, u₁(n) and u_(Q)(n), respectively, are produceddigitally in software and/or hardware using the digital signal processor(DSP) 1100. The baseband signals 1210 and 1220 are inputs to thedigital-to-analog converters (DACs) 1310 and 1320 respectively, whichproduce baseband analog output signals. The baseband in-phase andquadrature-phase signals are passed through low pass filters (LPF) 1312and 1322, respectively, which perform anti-image filtering. These arealso sometimes referred to as analog reconstruction filters by thoseskilled in the art. Filters 1312 and 1322 may also include gain,impedance conversion, and other signal conditioning functions. Thefiltered baseband signals are inputs to quadrature modulator 1380, whichalso gets local oscillator input from the RF synthesizer 1810, whichprovides the RF carrier signal to both the transmitter and receiver1900. The output signal from quadrature modulator 1380 passes into theRF power amplifier (RFPA) 1390 which produces a high power version to besent to one or more antennas 2000. Note that RFPA 1390 may consist ofone or more preamplifier and driver stages as well as the final outputdevice(s). In conventional amplitude modulated RFID transmitters theRFPA operates in a linear or quasi-linear mode such as class-AB.Practical efficiency for these types of amplifiers is in the 20% to 40%range in backscatter RFID applications given the peak-to-average ratioof the signals and linearity requirements involved.

Continuing with FIG. 2, the high power output of the RFPA 1390 isapplied to the TX-RX coupler 1820. The principal function of the coupler1820 is to send substantially most or all of the high power RFPA outputsignal to the antennas, while any signal coming into the reader from theantennas is mostly or all passed into the receiver 1900. The coupler1820 minimizes the transmitter reflections and leakage passed into thereceiver 1900. In conventional passive RFID reader design the coupler1820 is typically an active coupler system with multiple control signalsfrom the DSP used to adaptively null or cancel transmitter reflectionsand leakage into the receiver. An example of such a TX-RX coupler systemis in U.S. Pat. No. 7,327,802 B2, which employs a magnetic or ferritecirculator to minimize losses in the receiver path together with a 3 dBcombiner and various other signal tuning elements. Other embodiments ofcoupler 1820 may include vector modulators, PIN diodes, varactor diodes,or other active elements to adapt, control, tune, and cancel thetransmitter leakage into the receiver. As will be described more fullywith respect to FIG. 8 and FIG. 9, a useful embodiment for coupler 1820is a four port passive coupler similar to that described in Kim et al in“A Passive Circulator for RFID Application with High Isolation using aDirectional Coupler”, in Proceedings of the 36^(th) European MicrowaveConference, 2006. In practice, a reflective modulator must be attachedto the fourth port of the directional coupler to adaptively cancel thetransmitter leakage into the receiver. Thus, coupler 1820 may be any ofa number of adaptive systems which are commonly known to those skilledin the art to reduce transmitter leakage into the receiver.

The receiver 1900 uses a low noise amplifier (LNA) 1910 to add gain inthe RF analog stage prior to the down conversion mixer 1921, providingthe means to receive RFID tag responses. This is done is to improve thenoise figure of the receiver 1900. This will be discussed in greaterdetail with regard to FIG. 8 and FIG. 9. Continuing with FIG. 2, thereceive mixer 1921 performs a quadrature demodulation of the RF receivesignal to baseband using the local oscillator signal provided by RFfrequency synthesizer 1810. The baseband receive in-phase and quadraturephase signals are amplified by gain stages 1941 and 1951, respectively,and filtered by analog receive filters 1961 and 1971, respectively. Insome implementations there may be more than one gain stage and thefiltering may be distributed between the gain stages. In someimplementations the gain and filtering can be combined by using activefilters. The output of the baseband filtering and gain are input to thein-phase and quadrature-phase analog-to-digital (ADC) converters 1981and 1991, respectively. The ADCs convert the analog signals to digitalsignals for further processing and decoding within the DSP 1100. The DSP1100 interfaces with a client device to report tag responses.

FIG. 3 illustrates an architecture for RFID reader 1000 based on a polarmodulation transmitter 1400 and a low cost receiver 1901 with an activemixer 1922. In this design the DSP 1101 produces sampled data signalse(n) 1212 which represents the envelope of the RF signal and p(n) 1222which represents the phase of the RF signal. Note that the envelope andphase signals 1212 and 1222, respectively, may represent predistortedversions of the actual envelope and phase to compensate for linear andnonlinear distortion in the analog circuitry. The envelope 1212 andphase 1222 signals are inputs to the transmitter DACs 1410 and 1420,respectively, which are followed by low pass filters 1412 and 1422,respectively. The output of the LPF 1412 in the envelope amplifier inputsignal path is input to an envelope amplifier 1460. The envelopeamplifier is a fast tracking power converter or may accurately bereferred to as a modulated power supply. The envelope amplifier 1460provides a means to amplify the envelope amplifier input signal andincludes and/or is supplied by one or more fixed power supplies, whichis not shown in the block diagram of FIG. 3, and produces a high outputpower variable voltage which accurately tracks the input signal from theenvelope DAC 1410 and low pass filter 1412. The high power envelopeoutput of envelope amplifier 1460 is used as the power supply for aswitch mode RFPA 1490. The RFPA 1490 can be any type of saturated,switch mode RFPA such as class D, E, inverse E, F, inverse F, orsimilar, and provides a means to supply the transmit signal to thecoupler and ultimately to the antenna. It may be referred to hereinsimply as a “switch mode power amplifier.” Class E amplifiers are gooddesign choices for UHF and the lower microwave band, with theoreticalefficiency over 90%, although practical efficiencies are in the 60% to80% range due to switching and passive component losses, among otherfactors.

By a switch mode power amplifier, what is meant is a circuit designed toconvert DC power or low frequency power as from an envelope amplifierinto RF power by approximating an ideal switch operation, e.g., byswitching on and off at the radio frequency. The on/off switching occursat the desired radio frequency thereby converting low frequency power toRF power. An ideal switch has either zero volts across it while currentis flowing, or no current through it while it is off with voltage acrossit. Therefore, an ideal switch dissipates no power internally becausethere is never current flowing through it while there is voltage acrossit. Power devices such as RF MOSFET or RF BJT transistors canapproximate switches by operating in or near device saturation and/ordevice cutoff. Sometimes this document refers to this as a saturatedRFPA. This operation is distinctly different from the quasi-linear RFPAoperation used in conventional RFID transmitter design, most often classAB.

The total efficiency of the transmitter 1400 may be characterized by theproduct of the individual efficiencies for the envelope amplifier 1460and the RFPA 1490. For example, if the envelope amplifier converts thefixed power supply voltage(s) to a high power envelope signal with 80%efficiency, and the RFPA converts the time varying envelope amplifieroutput to RF energy with 80% efficiency, then the overall powerefficiency of the transmitter may be said to be 64% efficient. Given thecombined effect of both components on overall power efficiency, thedesign of both components 1460 and 1490 can be important. As discussedin the previous paragraph, a class-E amplifier may be a good choice for1490. Regarding the envelope amplifier, while it could use a linearregulator type supply to track the input reference from LPF 1412, thiswould be very inefficient. The envelope amplifier should use some formof switched mode power conversion to achieve efficiency acceptable for areduced cost RFID reader.

There are challenges in the design of a switch mode envelope amplifier.The main challenge is that the switching process produces noise whichmust be removed from the high power envelope output. The envelopeamplifier switching frequency should be at least several times higherthan the envelope bandwidth so that the switching noise can be filteredout. Higher switching frequencies mean the image and harmonic noise dueto the switching process are at higher frequencies and will require aless complex output filter to adequately remove them. The efficiency ofthe switch mode power supply depends on the switching frequency of theconverter with power efficiency of switching power supplies typicallydecreases as the switching frequency increases due to switching losses.These facts lead to us to an engineering design tradeoff in theswitching frequency parameter between efficiency and output filtercomplexity/output noise level. For low bandwidth signals such as thoseused in backscatter RFID systems a buck converter may be a good choicefor envelope amplifier 1460 due to straightforward circuitry and goodefficiency. Other topologies are possible for 1460, such as boost/buck,multilevel converters, multi-input converters, multi-phase converters,or hybrids of these including linear regulator assistance.

A switch mode power supply should be a central component of the envelopeamplifier since linear power supplies on their own have poor powerefficiency. Some embodiments of the envelope amplifier may combine alinear regulator in series or parallel with the switch mode supply. Thiswill be discussed in more detail with respect to FIG. 6 and FIG. 7. Someembodiments of the envelope amplifier will include explicit switchingcontrol from the DSP, as in FIGS. 5, 6, and 7. The embodiment of FIG. 3has only an analog envelope reference signal as input from LPF 1412. Inthis case the switch mode control will use hysteretic feedback control,V2 control, or some other form of switching power supply controller usedto track the reference analog input.

Continuing with the transmitter 1400 of FIG. 3, the phase signal 1222passes through DAC 1420 and low pass filter 1422 to be used as a controlsignal for phase shifter 1450. Phase shifter 1450 takes its input fromthe RF frequency synthesizer 1810 and serves as a phase modulator,providing a means to phase modulate the RFPA by shifting the phase ofthe signal provided. The phase shifter 1450 passes the local oscillatorsignal to the input of the RFPA 1490 after imparting a variable phaseshift on it, the phase shift value being a function of the controlsignal received from the phase signal 1222. Design of such variablephase shifters based on RF diode networks is known to those skilled inthe art. The input to the switch mode RFPA 1490 is typically considereda “constant envelope” signal, although some phase-to-amplitudemodulation may occur within the phase shifter 1450 due to phasedependent insertion loss of the phase shift circuitry. Given that theRFPA 1490 is a switch mode amplifier, small variations in the RFPA inputamplitude are not important since the driver circuitry for the RFPA 1490are typically nonlinear and are essentially just turning the main RFPApower device on and off. The high power output of the RFPA 1490 isfiltered to remove harmonics, which is a common operation with RFPAs andnot shown in FIG. 3 for clarity. The output of transmitter 1400 ispassed to the coupler 1820 which operates similar to that of FIG. 2.

Continuing with FIG. 3, the receiver output of the coupler 1820 ispassed into receiver 1901 where it is passively connected to the activemixer 1922. By passively connected it is meant that the signal is passedby means that include no active, or powered, circuit stages between theoutput of the coupler 1820 and the input to the active mixer 1922. Theactive mixer 1922 uses a local oscillator signal from RF frequencysynthesizer 1810 as input for the down conversion. By an active mixer itis meant a mixer which has a conversion gain instead of a conversionloss. The net gain of an active mixer 1922 would typically be around 3dB, depending on the baseband output impedance of the mixer and correctdesign of the matching circuitry at the output of the mixer 1922. Thereader design as shown in FIG. 3 is optimized for passive backscatterRFID. By using an active mixer 1922 which is passively connected to thecoupler 1820, the noise figure of the RFID receiver is sufficient forpassive backscatter RFID according to the ISO 18000-6 and Gen2protocols, while maintaining very high linearity, low cost, and lowpower consumption. Conventional passive RFID reader design for ISO18000-63 and Gen2 protocols use at least one LNA as in receiver 1900.Some reader designs even use more than one LNA in the receiver. However,FIG. 8 and FIG. 9 of this specification will disclose why this is aninferior architecture to the low RF gain architecture of receiver 1901illustrated in FIG. 3. Receiver 1901 is optimized for performance andcost in RFID systems using Gen2 and/or passive ISO 18000-63.

Continuing with FIG. 3, the in-phase and quadrature-phase basebandoutputs of the active mixer 1922 are passed to gain stages 1942 and1952, respectively, then to baseband filters 1962 and 1972,respectively, and finally into the analog-to-digital converters (ADCs)1980 and 1990, respectively. The outputs of the ADCs 1980 and 1990 arepassed into the DSP 1101 for decoding tag responses, which are thenpassed from the DSP 1101 to the client device. Note that the basebandlineup shown in FIG. 3 is just one example embodiment. The filtering maybe interspersed with the gain, or the filtering may be combined with thegain by using active filters. The gain stages 1942 and 1952 willcommonly occur near the output of the active mixer 1922 so that thenoise figure can be maintained. However, some designs may use an ACcoupled receiver topology for which a DC blocking high pass filtersection may be employed between the active mixer 1922 and the gainstages 1942 and 1952. Blocking the DC signal content before the gain isoften advantageous since the residual carrier leakage in the homodynesystems shown in the example embodiments of this disclosure can be veryhigh and can cause compression and linearity problems if it is allowedto pass through the gain stages. The gain stages 1942 and 1952 mayemploy transformers or autotransformers as components to simultaneouslysupply voltage gain, impedance matching, and high pass filtering.

Polar transmitters such as in FIG. 3 may operate by separating theenvelope and the phase of the baseband transmit signal using atransformation similar to

e(n)=√{square root over (u _(I)(n)² +u _(Q)(n)²)},

and

p(n)=a tan(u _(Q)(n), u _(I)(n)),

for the envelope and phase, where u_(I)(n) and u_(Q)(n) are theCartesian in-phase 1210 and quadrature-phase 1220 baseband signals ofFIG. 2, which together form the complex baseband signal

u(n)=u _(I)(n)+ju _(Q)(n),

where j=√{square root over (−1)}. This nonlinear transformation fromCartesian to polar coordinates may be performed numerically inside DSP1101 prior to sending the envelope amplifier input signal to the DACs1410 and 1420. Alternatively, the envelope and phase signals may begenerated using table driven signal synthesis as disclosed in U.S. Pat.No. 9,813,115 B2 entitled, “RF System Using PR-ASK with OrthogonalOffset,” incorporated herein by reference. The polar transmitter workson the polar signal representation

u(n)=e(n)·exp(jp(n))

essentially by power amplifying the envelope amplifier input signal andapplying the high power envelope signal to a constant envelope RF signalsource which has had the phase signal p(n) modulated onto it. The highpower output of the polar transmitter 1400 is

u _(out)(t)=e _(out)(t)·exp (j(ωt+p(t))),

which is a continuous time RF passband version of the original basebandsignal u(n). Other approximations and implementations of the nonlineartransformation to get the envelope and phase are possible and do notdepart from the ideas disclosed herein. Indeed, it is possible todirectly create the waveform in the polar coordinate system without thetransformation above, yet this is still within the scope of the RFIDpolar transmitter concepts disclosed herein.

While there are significant benefits to the polar transmitterarchitecture for RFID readers which are enumerated throughout thisdisclosure, there are also a number of challenges, including:

-   -   Bandwidth expansion in the envelope e(n) and phase p(n) signals        resulting from the implicit nonlinear transformations used to        get these signals    -   Time alignment between the envelope and phase paths in the polar        transmitter must be carefully matched    -   Deep amplitude modulation, such as amplitude modulation at or        near 100%, is typically not feasible because the switch mode        RFPA 1490 requires some minimum operating voltage with which to        function properly        The bandwidth expansion is an important but subtle concept which        will be described further now with reference to FIG. 4. The        bandwidth of the baseband signal u(n) and the transmitter output        signal u_(out)(t) are ideally the same, with the output signal        simply translated up to RF frequency f=ω/2π. In practice the        power spectrum of u_(out)(t) will have some additional “spectral        regrowth” artifacts due to nonlinear distortion in the        transmitter. The two component signals of the polar        representation e(n) and p(n) individually will often have wider        bandwidths due to the implicit or explicit nonlinear        transformation used to generate these components, even though        when recombined using u(n)=e(n)·exp(jp(n)) they result in the        original signal bandwidth. The bandwidth expansion of the        envelope amplifier input signal e(n) is particularly problematic        because of tradeoffs between bandwidth and power efficiency in        the envelope amplifier 1460.

For passive and semi-passive backscatter RFID systems the most commonreader transmit data encoding and modulation formats in commercial useare

-   -   Manchester encoded AM,    -   PIE encoded large carrier AM (referred to as DSB-ASK), and    -   PIE encoded suppressed carrier AM (referred to as PR-ASK).        Of these formats the Manchester and DSB-ASK may have no envelope        bandwidth expansion since it may be possible, depending on the        modulation depth, to express the envelope as e(n)=u_(I)(n) for        these transmit signals. However, Manchester and DSB-ASK are        significantly less bandwidth efficient than PR-ASK. Bandwidth        efficiency is typically defined in terms of bits per second per        hertz, in other words, the data rate achieved per unit        bandwidth. Thus, even though Manchester or DSB-ASK modes have        little or no bandwidth expansion in their envelope amplifier        input signal, the PR-ASK signal format is often preferred since        the actual transmission signal achieves a higher bandwidth        efficiency even though its envelope amplifier input signal has        significant bandwidth expansion.

In addition to the envelope bandwidth expansion, there are othersignificant obstacles to implementing a polar transmitter using thePR-ASK mode, specifically:

-   -   The fully modulated, theoretical 100% amplitude modulation depth        is not practical to implement due to minimum operating voltage        requirements of the switch mode RFPA output device        -   This applies to Manchester and DSB-ASK as well if they are            fully modulated, i.e., at or near 100% amplitude modulation            depth    -   The polarity inversions in PR-ASK lead to discontinuities in the        phase signal p(n) 1222 which are not possible to reproduce at        the input to the phase shifter 1450 or within the phase shifter        circuitry    -   The time alignment between the phase and envelope signals as        they effect the RF output signal require such high precision due        to the abrupt, near 180 degree phase inversions in conventional        PR-ASK        A modulation format referred to as OPR-ASK is introduced in the        previously mentioned U.S. Pat. No. 9,813,115 B2 entitled, “RF        System Using PR-ASK with Orthogonal Offset.” The OPR-ASK reader        transmission format achieves a slightly better bandwidth        efficiency than PR-ASK but is readily implemented using a polar        transmitter since it does not suffer the obstacles pointed out        above.

FIG. 4 contains plot 5000 of the power spectrum for several examplessignals. The power spectra shown are for continuously modulated signalswith random data. The power spectra for OPR-ASK 5100 and PR-ASK 5200 areshown to lie almost on top of one another. The only exception is near 0hertz (DC) where the OPR-ASK spectra 5100 rises up several dB above thePR-ASK spectrum 5200. This is because the orthogonal offset in theOPR-ASK modulation technology creates a small DC component. The powerspectrum 5300 is for the envelope amplifier input signal e(n) 1212 whenthe RFID reader transmit signal modulation is OPR-ASK, while the powerspectrum 5400 is for PR-ASK modulation. These power spectra are shownwith the average DC power supply component removed for clarity, so onlythe spectrum of the time varying envelope component is shown. The PR-ASKsignal has frequent zero crossings in the baseband signal which causessignificant bandwidth expansion in the envelope amplifier input signale(n) 1212. The bandwidth of the envelope amplifier input signal is animportant factor since this may affect the switching frequency andefficiency of the envelop amplifier.

An example embodiment of the RFID polar transmitter specified hereinemployed OPR-ASK modulation because of the following very importantqualities:

-   -   Reduced bandwidth expansion of the OPR-ASK envelope amplifier        input signal    -   Reduced amplitude modulation depth of OPR-ASK which allows        practical implementation of the RFPA because the supply voltage        never goes to zero volts    -   A continuous and continuously differentiable OPR-ASK phase        signal which allows practical implementation of the phase        modulator, phase shifter, or phase controller    -   The smooth, continuously differentiable nature of the OPR-ASK        envelope and phase signals allows easier time alignment between        the envelope and phase components at the RFPA device, which in        turn yields reduced distortion output        The OPR-ASK modulation format facilitates a more practical        realization of the RFID polar transmitter.

FIG. 5 illustrates another example embodiment of this disclosure basedon a polar modulation transmitter 1500. In this embodiment of the polarmodulation transmitter 1500, the DSP 1102 produces a pulse widthmodulated (PWM) signal pwm(t) 1214 which represents the envelope of theRF signal. The PWM signal may still be referred to herein as the“envelope amplifier input signal.” The PWM signal 1214 is the input toan envelope amplifier 1560 which converts the PWM signal 1214 to a highpower envelope output signal which varies in proportion to the dutycycle of the PWM signal. The example embodiment of envelope amplifier1560 is a buck converter, which is straightforward and has highefficiency. In order to improve efficiency the buck converter ofenvelope amplifier 1560 employs a second MOSFET 1565 in place of a catchdiode. This technique is referred to as synchronous rectification. ThePWM signal 1214 is input to a switch logic component 1561 whichimplements dead-time controller to avoid power supply shoot through inthe synchronous rectifier, a technique which is understood by switchmode power supply engineers. Based on the input PWM signal, the switchlogic controller controls the switch driver 1562 for the top switch 1564and also controls the switch driver 1563 for the bottom switch 1565. Thebuck converter as illustrated in FIG. 5 has an output filter made up ofinductors 1566 and 1568 and capacitors 1567 and 1569. This example is a4^(th) order output filter, but other orders are possible. The frequencyresponse of the output filter could be Bessel, Butterworth, Legendre, orsome other type of response. The response must attenuate the PWMswitching harmonics and mixing images without excessive phase oramplitude distortion of the baseband envelope signal. If the DC powersupply to the buck converter is constant and the buck converter operatesin continuous conduction mode, then the envelope amplifier 1560 haslinear output proportional to the PWM input duty cycle. This embodimenteliminates the DAC 1410 and filter 1412 associated with polar modulationtransmitter 1400 in FIG. 3. Many common DSPs have PWM output capability.One example is the F28xxx family of DSPs produced by Texas Instrumentswith integrated high resolution PWM (HRPWM), which may achieve betterthan 12 bits resolution with a switching frequency of 1.25MHz. Fullycustom DSP solutions for RFID polar modulation may achieve even higherperformance. The analysis and design of output filters for buckconverters to suppress the switching frequency, harmonics, and sidebandsis understood by those skilled in the art.

Continuing with FIG. 5, the DSP 1102 also produces a phase signal p(n)1224 which is input to a direct digital synthesizer (DDS) 1550, whichserves as a phase modulator. The DDS 1550 is capable of synthesizing RFsignals in the appropriate UHF or microwave frequency band. The DDS 1550can also digitally phase modulate the RFPA using the phase signal 1224from the DSP 1102. This embodiment eliminates the DAC 1420, filter 1422,analog phase shifter 1450, and RF frequency synthesizer 1810 associatedwith the RFID reader design of FIG. 3. The output of DDS 1550 is asubstantially constant envelope phase modulated signal which is input tothe switch mode RFPA 1490, whose power supply is envelope modulated byenvelope amplifier 1560. The output of DDS 1550 is also passed to thereceiver 1901 for use in down-converting the receive RF signals tobaseband for processing. The remaining operation of the coupler 1820 andreceiver 1901 is the same as for the embodiment of FIG. 3.

FIG. 6 illustrates another example embodiment of this disclosure basedon a polar modulation transmitter 1600. In this embodiment of the polarmodulation transmitter 1600, the DSP 1103 produces both a PWM signalpwm(t) 1214 and a signal e(n) 1212 which represents the envelope of theRF signal. The two of these signals together may be referred to hereinat times as the “envelope amplifier input signal.” DAC 1410 and LPF 1412are used to create a continuous version of the envelope e(t), which isused as a reference signal in the linear assisted switch mode envelopeamplifier 1660. The principal reason for using the linear assistance isfor better tracking of the envelope amplifier input signal since alinear regulator will be able to provide a higher slew rate andcompensate for tracking errors which may occur with a switch modeenvelope amplifier alone. There are two basic choices of how to combinethe switch mode supply with the linear regulator supply: series orparallel. In series operation the switch mode power supply provides thetime varying operating voltage from which the linear regulator operates.The linear regulator provides the fast tracking output voltage, and thedesign goal of the series system is to minimize the voltage drop acrossthe linear regulator. In a parallel combination of the switcher andlinear amplifiers the linear regulator provides error correction to theswitcher output. The envelope amplifier 1660 of FIG. 6 illustrates thearchitecture for a series linear regulator. FIG. 7 will provide adetailed example of parallel operation.

The envelope amplifier 1660 uses a switch mode power converter 1661 toproduce a time varying output supply voltage. The converter 1661 couldbe a buck converter similar to in FIG. 5, or it could be a boost-buckconverter, a multi-level converter, or any other switch mode supplyknown to those skilled in the art. The difference as applied to theseries linear regulator combination is that the time varying output mustbe a slightly higher voltage than the actual desired envelope amplifieroutput. The linear pass element 1662 drops the output voltage to thedesired output value. The pass element is controlled by error amplifier1663 which accepts the LPF 1412 output as the reference and senses theactual output using feedback resistors 1664 and 1665. The linearregulator consisting of 1662, 1663, 1664, and 1665 is a low noise, highbandwidth system able to accurately track the reference signal. There istypically some minimum voltage drop across the pass element 1662, andthe efficiency of this series assisted envelope amplifier is optimizedwhen the switch mode converter 1661 keeps its output voltage near thedesired output plus the minimum voltage drop. The pass element 1662 isthe power stage of the linear regulator and may be one of a variety oftechnologies such as a NPN device, PNP device, NMOS device, or PMOSdevice. Note that the envelope reference signal 1212 must be delaymatched with the PWM signal 1214 for optimum performance. This isillustrated in detail in FIG. 7. The remaining operation of the systemin FIG. 6 is the same as in FIG. 3.

FIG. 7 provides a detailed example embodiment of polar transmitter 1700.This embodiment uses a linear assisted switch mode power supply wherethe linear regulator operates in parallel with the switch mode powersupply. Note that the components 1120, 1130, 1140, 1150, and 6401 areall part of the DSP 1103. The demux 6401 is part of a table drivenapproach as in U.S. Pat. No. 9,813,115, except in this embodiment thetable waveforms contain four channels of data representing the waveformspwm₊(n) 1213, pwm⁻(n) 1215, e′ (n) 1211, and p′ (n) 1221. The demux 6401separates the data channels of the stored waveform as they are read frommemory and sends each channel to its designated output. The pwm₊(n) andpwm⁻(n) outputs are sent to PWM timers 1120 and 1130, respectively.These timers produce continuous time PWM signals pwm₊(t) 1214 andpwm⁻(t) 1216 which are used to control the main power switches 1761 and1762 within the envelope amplifier 1760. Note that having two separatePWM outputs as in FIG. 7 eliminates the need for switch logic 1561 andin FIG. 5. The output of the switching devices passes through a filter1764 which filters the switching waveforms before connecting to thecombiner 1765 where the switching and linear outputs are combined.

The envelope amplifier input signal 1211 from the demux 6401 is passedto a delay 1140 which is optimized to minimize the delay mismatchbetween the PWM path to the combiner and the linear regulator path tothe combiner. The output of delay 1140 goes to the DAC 1410 whichconverts the sampled data signal to a continuous time signal. The DAC1410 will typically be followed by LPF 1412. The continuous timeenvelope amplifier input signal passes to the drive amplifier 1763 whichprovides level and impedance conversion for input to the linearregulator 1766. The linear regulator would typically be a current modeamplifier which compares the input from driver 1763 with the outputvoltage v_(out)(t) using feedback resistors 1767 and 1768. The output ofthe linear regulator is combined with the switching converter atcombiner 1765. The circuitry for the combiner 1765 make take on severalforms, such as a push-pull stage with a pair of NPN and PNP bipolartransistors or it could be a pair of diodes connected anti-parallel,Schottky diodes being best due their low forward voltage drop. Thepurpose of the parallel linear regulator is to reduce the error in theoutput voltage. The switching converter efficiently provides the bulk ofthe output power but may not be able to provide a precisely trackingoutput voltage. The linear regulator is less efficient but only needs todrive small corrections in the output voltage to accurately track theinput. Note again the fundamental difference between the series andparallel combination of the switching converter and linear regulator: ina series architecture the linear regulator operates continuously off theswitcher output to track the reference signal, while in the parallelarchitecture the linear regulator operates off a fixed supply voltagebut only sources or sinks current to make corrections to the switcheroutput.

The phase signal 1221 from the demux 6401 is passed to a delay 1150which is optimized to minimize the delay mismatch between outputs of thephase shifter 1750 and the envelope amplifier 1760. The output of delay1150 goes to the DAC 1420 which converts the sampled data signal to acontinuous time signal. The DAC 1420 will typically be followed by LPF1422. The continuous time phase control signal goes to driver amplifier1752 which controls a diode network 1756 configured to providecontinuous phase shifting of the RF signal under the control of driver1752. The diode network is best designed using varactor diodes but couldbe done with pin diodes as well. Other phase shift circuits are possibleand do not depart from the ideas disclosed herein. The design of acontinuously variable phase shift network using RF diodes is known tothose skilled in the art. The output of the frequency source 1810 passesto isolation amplifier 1754. The diode network 1756 typically will havea high insertion loss as well as an input impedance which varies basedon the control signal from driver 1752. The isolation amplifier 1754 maybe needed to isolate the frequency source 1810 from the time varyingreturn loss of the diode network. The isolation amplifier can alsoprovide signal gain to fulfill to total gain requirement of thetransmitter lineup. The output of the diode network 1756 is passed tothe amplifier 1770. The amplifier 1770 provides gain and matching todrive the main RF output switch mode device in the RFPA 1790. The RFPA1790 converts the time varying envelope voltage from the envelopeamplifier to an RF output signal by essentially being switched on andoff with the phase modulated RF signal from driving amplifier 1770.

Beyond power efficiency, there are additional commercial advantages ofusing a switch mode RFPA such as a class-E amplifier. First, since theRFPA is operating in a fully saturated mode the broadband output noisefloor of the transmitter may be reduced since the mechanism of gain andnoise figure do not apply the same way as for linear mode devices. Thisreduces amplitude noise but does not affect phase noise, which mostlyarises due to the synthesizer signal driving the RFPA. A secondadvantage is that the switch mode amplifier may be more readilyintegrated onto a system-on-a-chip than conventional class-AB designs. Athird advantage is that switch mode designs such as class E may havereduced sensitivity to varying output load impedance, which can lead tomore robust performance.

FIG. 8 displays an example link budget for typical modern RFID systemswere tag sensitivities are in the range of −10 dBm to −25 dBm. Inpassive RFID systems the tags 4000 rely on the forward radiating powerfrom the RFID reader's transmitter 8100 to power the tags. For thepurposes of the link budget analysis in FIG. 8 the reader transmitter8100 may be any of transmitters 1300, 1400, 1500, 1600, 1700, or somecombination of these. Under most regulatory environments the transmitter8100 can conduct approximately one watt, or +30 dBm into the antenna.The forward path loss is given by the Friis equation, well known tothose skilled in the art. The best tag sensitivity as of this writing isin the range of −18 dBm. This is the required power harvested at the tag4000 for the tag to power up and activate. With future improvements insemiconductor technology tags may achieve −20 dBm or even better. Using−20 dBm as the minimum tag sensitivity, this suggests the range, andtherefore forward path loss, is limited to 50 dB.

Not all of the power reaching the tag is modulated back to the reader.This reduction in backscatter modulation level is referred to asconversion loss in FIG. 8. A conversion loss of 5 dB is typical.Referring to FIG. 8, if the reader's transmit signal reaches the tag ata received power level of −20 dBm, then for a 5 dB backscatterconversion loss the tag signal sent back to the reader is −25 dBm. Formonostatic antenna RFID systems the radio path is reciprocal, meaningthe reverse path loss is the same as the forward path loss. Given the−20 dBm minimum activation power, the 5 dB conversion loss, and the 50dB reverse path loss, it can be seen that the minimum receive signalstrength at the reader will be −75 dBm.

For the common signal encoding used in RFID backscatter modulation asignal-to-noise ratio (SNR) of from 10 dB to 15 dB is required toachieve desired performance levels. FIG. 8 uses 15 dB as the SNRrequirement, which yields a maximum noise floor from the receiver 8200of −90 dBm. For the purposes of the link budget analysis in FIG. 8 thereader receiver 8200 may be any of receiver 1900, 1901, or somecombination of these. In the absence of self-noise from the reader's owntransmitter, it is well known to those skilled in the art that the radioreceiver will have a minimum noise set by the thermal noise in thereceiver together with the receiver's bandwidth BW and noise figure NF.The thermal noise density N₀, in watts per hertz, is

N ₀ =k·T,

where k is Boltzmann's constant and Tis the receiver temperature inKelvin. This corresponds to approximately −174 dBm/Hz at 290 degreesKelvin. The total thermal noise power into the receiver depends on thereceiver bandwidth in Hertz. Using 1 MHz receiver bandwidth correspondsto 60 dB, i.e., this is

BW=10·log₁₀(10⁶ Hertz)=60 dB,

then the total thermal noise power into the receiver is k·T·BW=−114 dBm.Note that there are many ways to measure the bandwidth of a system.Those skilled in the art are familiar with the concept of equivalentnoise bandwidth, which is the definition used herein. The −114 dBm noiselevel would suggest a 24 dB margin compared to the −90 dBm maximumreceiver noise output calculated above and shown in FIG. 8 to achievethe desired performance level. However, all receivers add a certainamount of noise to the theoretical minimum. This added noise above thetheoretical minimum is referred to as noise figure NF, measured in dB.

Those skilled in the art know how to calculate the noise figure of areceiver from the line-up of components that make up the receiver. FIG.9 shows a generic component lineup for an RFID receiver. The receivermay consist of circulators, isolators, directional couplers, low noiseamplifiers, mixers, baseband amplifiers, limiting diodes, as well asother possible components. Each individual component is characterized bya gain (or loss) and noise figure. Components which attenuate thesignal, i.e., components with a loss L have a noise figure equal to theloss NF=L in dB. Active components which have gain are characterized bytheir gain G and by their noise figure NF. The overall receiver noisefigure can be calculated using the Friis formula for a cascade ofcomponents. The Friis formula uses the linear gain g_(n) and noisefactor F_(n) of the n^(th) component in the line-up, where

F_(n)10^(NF) ^(n) ^(/10),

is the relationship between the linear noise factor F_(n) and the noisefigure NF_(n) in dB of the n^(th) component, and

g_(n)=10^(G) ^(n) ^(/10),

is the relationship between the linear gain g_(n) and the gain G_(n) indB of the n^(th) component.

For N components in the cascaded line up, where N=4 in FIG. 9, the totalnoise factor can be found with the Friis formula

$F = {F_{1} + \frac{F_{2} - 1}{g_{1}} + \frac{F_{3} - 1}{g_{1} \cdot g_{2}} + \frac{F_{4} - 1}{g_{1} \cdot g_{2} \cdot g_{3}} + \ldots + \frac{F_{N} - 1}{\prod\limits_{n = 1}^{N - 1}\; g_{n}}}$

The total noise figure of the receiver is NF=10·log₁₀(F).

The conventional RFID reader receiver 1900 of FIG. 2 uses an LNA 1910for a gain stage at RF in order to essentially set the noise figureahead of the demodulation mixer 1921. Typical demodulation mixers haveconversion loss from 9 dB to 13 dB, with the mixer noise figure equal toits loss, i.e. NF=L. High quality LNAs typically have 10 dB to 20 dB ofgain with a noise figure of 2 to 4 dB. The baseband gains and filters inRFID receivers 1900 and 1901 each have associated gains (or losses) andnoise figures.

The table below shows the gain-noise figure analysis for theconventional RFID receiver 1900 line up shown in FIG. 2.

Com- Com- Com- Com- Com- ponent 1 ponent 2 ponent 3 ponent 4 ponent 5Gain “G” (dB) −0.5 −3.0 17.0 -9.0 20.0 Gain “g”(linear) 0.9 0.5 50.1 0.1100.0 Noise Figure, “NF” (dB) 0.5 3.0 2.5 9.0 15.0 Noise Factor, “F”(linear) 1.1 2.0 1.8 7.9 31.6 Cumulative gain, g (linear) 0.9 0.4 22.42.8 281.8 Cumulative gain, G (dB) −0.5 −3.5 13.5 4.5 24.5 CumulativeNoise Factor 1.1 2.2 4.0 4.3 15.2 Cumulative Noise Figure (dB) 0.5 3.56.0 6.3 11.8 TX-RX Active Receiver Receiver Baseband Ferrite TX-RX LNAMixer Gain & Circulator Cancellation Filter SummingIn this implementation First component 1 (referring to FIG. 9, the tableabove, and the Friis noise figure formula) is a ferrite circulatorwithin coupler 1820 which has 0.5 dB of loss. Second component 2 is atransmitter cancellation RF signal combiner with a 3 dB loss. Thissecond component 2 is also part of the coupler 1820. Third component 3is a low noise amplifier 1910 with a gain of 17 dB and noise figure of2.5 dB. These are typical numbers for high linearity LNAs in the UHF andmicrowave bands. Forth component 4 is quadrature down conversion mixer1921 with a conversion loss of 9 dB. In this case the receiver lineuphas a fifth component comprising the baseband amplifier and filteringwith 15 dB noise figure and 20 dB of gain. Baseband amplifiers generallyhave much poorer noise figure than RF amplifiers. The gain and noisefigure calculations shown in the table above result in a final gain andnoise figure being G=24.5 dB and NF=11.8 dB. This is clearly better thanthe maximum noise figure of 24 dB shown in the example link budget ofFIG. 8. It is often said that in a well-designed receive chain only thenoise factor of the first amplifier should be significant. The lineup inFIG. 2 and the table above is an example of this design mindset.However, the design of FIG. 2 is expensive and may suffer frominadequate linearity due to the higher broadband gain at RF. Thebaseband includes filtering to reduce the bandwidth down to the a fewkilohertz or up to a megahertz or so. This limits the interference andpreferably the receiver gain is placed here in baseband whereout-of-band interference is removed.

The table below shows the gain-noise figure analysis for the RFID readerdesign disclosed in FIG. 3.

Com- Com- Com- Com- ponent 1 ponent 2 ponent 3 ponent 4 Gain “G” (dB)−6.0 3.0 20.0 10.0 Gain “g”(linear) 0.3 2.0 100.0 10.0 Noise Figure,“NF” (dB) 6.0 12.0 15.0 15.0 Noise Factor, “F” (linear) 4.0 15.8 31.631.6 Cumulative gain, g (linear) 0.3 0.5 50.1 501.2 Cumulative gain, G(dB) −6.0 −3.0 17.0 27.0 Cumulative Noise Factor 4.0 63.1 124.2 124.8Cumulative Noise Figure (dB) 6.0 18.0 20.9 21.0 TX-RX Active BasebandBaseband Coupler Mixer Gain & Gain & Filter Filter (optional)In this implementation first component 1 is a directional coupler usedas a four port device for transmitter cancellation via the reflectivemodulator mechanism analogous to the approach described by Kim et al in“A Passive Circulator for RFID Application with High Isolation using aDirectional Coupler”, in Proceedings of the 36^(th) European MicrowaveConference, 2006. The directional coupler is part of the coupler 1820and has a receiver coupling loss of 6 dB in this example. The couplingfactor is a design decision which affects the receiver noise figure andthe post power amplifier loss in the transmission path. Higher couplingpresents less loss in the receiver path, but more loss in thetransmitter path. An example embodiment presented here opts for a 6 dBcoupling factor.

First component 1 is passively connected to Second component 2, which isan active quadrature down conversion mixer 1522 with a conversion gainof 3 dB and a noise figure of 12 dB. Third component 3 comprises thebaseband amplifier and filtering with 15 dB noise figure and 20 dB ofgain. The gain and noise figure calculations shown in the table aboveresult in a gain and noise figure being G=17.0 dB and NF=20.9 dB at theoutput of third component 3. If more gain is desired to reduce theeffects of quantization noise at the ADCs 1980 and 1990, then anoptional forth component 4 of additional gain and filtering can be addedwith 10 dB of gain and a noise figure of 15 dB. This puts the total gainand noise figure calculations being G=27.0 dB and NF=21.0 dB at theoutput of third component 4. This is better than the maximum noisefigure of 24 dB shown in the example link budget of FIG. 8, but atreduced cost, reduced complexity, and improved linearity. Indeed, whilethe very low noise figure design of FIG. 2 may be needed forsemi-passive backscatter tags which have much lower sensitivity than −20dBm, the design of FIG. 2 is excessively complex and costly for passivebackscatter RFID tags, as disclosed herein.

Note that elements of FIG. 3, FIG. 5, FIG. 6, and FIG. 7 can beinterchanged without affect in the novel elements disclosed herein. ThePWM driven envelope amplifier 1560 of FIG. 5 could be used with theanalog phase modulation design of FIG. 3, and conversely the DDS 1550 ofFIG. 5 could be used with the analog input envelope amplifier 1460 ofFIG. 3. Envelope amplifiers designed as linear assisted switch modeamplifiers can be used with any type of phase modulator. All of thesetransmitters and even transmitter 1300 can be used with a passivelycoupled receive RF path and active mixer 1922 in order to reducereceiver cost and improve linearity. Other variants are also possiblewithout departing from the ideas disclosed herein and will be evident tothose skilled in the art.

FIG. 10 shows a block diagram of an alternative embodiment for an RFIDsystem using a polar transmitter wherein the transmit and receivefunctions may be physically separated. RFID transmitter 1001 contains aDSP 1104 which performs baseband signal processing and radio controlfunctions for the transmitter. DSP 1104 sends envelope and phase signalsto polar transmitter 1601, which may be any combination of polartransmitter elements previously discussed. The high power polartransmitter output is send to one or more transmit antennas 2001connected to transmitter 1001. The transmitter 1001 may contain its ownfrequency source 1811 as shown in FIG. 10, or it could be feed afrequency source from some external master clock source which is notshown. The RFID receiver 1002 contains a receiver 1902 which isconnected to one or more receive antennas 2002. Receiver 1902 may besubstantially different from previously discussed receivers since thisis a bistatic antenna configuration and the noise, transmitterself-interference, and sensitivity requirements may be substantiallydifferent than for the monostatic configurations previously discussed.Recall that a monostatic antenna configuration is when the same antennais used for simultaneously transmitting and receiving. The RFID receiver1002 contains a DSP 1105 which demodulates tags. The RFID receiver mayhave its own frequency source 1812, may be feed from the RFIDtransmitter's frequency source 1811, or could be feed from some othermaster frequency source. Both RFID transmitter 1001 and RFID receiver1002 communicate to the client hardware or software. There will alsocommonly be timing coordination between the RFID transmitter 1001 andreceiver 1002 to control the protocol timing, among other things.

FIG. 11 shows a block diagram of an alternative embodiment for an RFIDpolar transmitter 1502. In this example, the PWM control signal 1217 isthe input to switch logic 1571. Switch logic 1571 controls four sets ofswitch drivers, 1572, 1573, 1574, and 1575. The switch drivers here aresimilar in function to 1562 and 1563. The difference is that the lefttop/bottom switch pairs 1576 and 1577 are driven 180 degrees out ofphase from the right top/bottom switch pair 1578 and 1579. This is atwo-phase buck converter which can be generalized to a multi-phase buckconverter. The left and right switch pairs are connected through amatched pair of inductors 1581 and 1582. In the example of FIG. 11, theparallel combination of inductors 1581 and 1582, together with shuntcapacitor 1583 and series inductor 1584 form a 3^(rd) order low passoutput filter for the envelope amplifier 1860.

The envelope amplifier 1860 in FIG. 11 provides supply modulation to themain RF power amplifier switch 1595 through RF choke 1593. In some RFswitches the bias and input drive level at the gate or base of thetransistor may need to be increased as the supply modulation signal goesthrough the low troughs of the envelope waveform. The example embodimentin FIG. 11 provides bias modulation 1218 through DAC 1591. The output ofDAC 1591 modulates the gate or base bias to provide the bias modulationthrough RF choke 1592. Likewise, drive level modulation is supplied bysignal 1219 into DAC 1597. The DAC 1597 output is the control input toprovide the drive level modulation to a voltage-controlled amplifier1598. Voltage-controlled amplifier 1598 varies the drive level into theRF power amplifier switch 1595 through a DC block capacitor 1594.

In the embodiment of FIG. 11, the phase signal 1225 is input to a phasemodulating digital frequency synthesizer 1551 which provides directphase modulation. Referring to the final components in FIG. 11, theoutput of RF switch 1595 passes through an output matching network 1596and is passed on the antenna & receiver coupler 1820.

FIG. 12 shows a block diagram of an alternative embodiment for an RFIDpolar transmitter 1503. In this example, the envelope amplifier 1960 iscontrolled using the PWM signal 1217. The envelope amplifier 1960 inFIG. 12 provides supply modulation to the main RF power amplifier switch1595 through RF choke 1593. In some RF switches the bias and input drivelevel at the gate or base of the transistor may need to be increased asthe supply modulation signal goes through the low troughs of theenvelope waveform. The example embodiment in FIG. 12 provides biasmodulation 1218 through DAC 1591. The output of DAC 1591 modulates thegate or base bias to provide the bias modulation through RF choke 1592.Likewise, drive level modulation may be needed to improve dynamic rangeor linearize the polar transmitter 1503. Using a quadrature modulator1381 and cartesian inputs 1231 and 1232 through DACs 1311 and 1321, thedirect conversion system using quadrature modulator 1381 and frequencysynthesizer 1813 provides both angle modulation and drive levelmodulation. Anti-imaging lowpass filters, not shown, can optionallyfollow the DACs in FIG. 12. Also, optional but shown in FIG. 12, is adriver amplifier 1599. The switch drive signal couples into the RF poweramplifier switch 1595 through a DC block capacitor 1594. This switchdrive signal has all the angle modulation for the polar transmitter sothat the frequency synthesizer 1813 can be a simple, unmodulatedcontinuous wave source. The frequency source 1813 also provides theclock reference to the RFID receiver 1902.

The example devices and methods in this disclosure can achieve FCC andETSI regulatory compliance with low cost, low power, and highperformance. In some embodiments, a general-purpose processor such as aDSP, microcontroller or microprocessor is used and firmware, software,or microcode can be stored in a non-transitory storage medium that isassociated with the device. Any such device may be referred to herein asa “processor” or a “microprocessor” and can execute the instructionsstored in the non-transitory storage medium. Such a medium may be amemory integrated into the processor, or may be a memory chip that isaddressed by the controller to perform control functions. Such firmware,software or microcode is executable by the processor and when executed,causes the processor to perform its control functions. Such firmware orsoftware could also be stored in or on a non-transitory medium such asan optical disk or traditional removable or fixed magnetic medium suchas a disk drive used to load the firmware or software into an RFIDsystem.

It should be noted that any data and information necessary to supportthe execution of instructions for any embodiment of the disclosure canbe placed in a removable storage medium as well. These could be storedon a disk as well, especially for development purposes or formaintenance and update purposes. Such a storage medium may be accessedeither directly or over a network, including the Internet.

Although specific embodiments have been illustrated and describedherein, those of ordinary skill in the art appreciate that anyarrangement which is calculated to achieve the same purpose may besubstituted for the specific embodiments shown and that the inventionhas other applications in other environments. This application isintended to cover any adaptations or variations of the presentinvention. The following claims are in no way intended to limit thescope of the invention to the specific embodiments described herein.

1. A radio frequency identification (RFID) system comprising: a receiverto receive responses from RFID tags; and a polar transmitter, the polartransmitter further comprising: a power amplifier to produce atransmitter output signal; an envelope amplifier connected to the poweramplifier to supply an envelope signal to the power amplifier; and aquadrature modulator connected to the power amplifier to providemodulation for the transmitter output signal using a cartesian inputsignal.
 2. The RFID system of claim 1 further comprising a couplerconnected to the polar transmitter and the receiver to pass thetransmitter output signal to one or more antennas and to pass theresponses to the receiver.
 3. The RFID system of claim 1 wherein thereceiver and the polar transmitter are separable.
 4. The RFID system ofclaim 1 wherein the envelope amplifier further comprises a multi-phasebuck converter.
 5. The RFID system of claim 1 further comprising: adriver amplifier connected to the power amplifier; and a DC blockingcapacitor connected between the driver amplifier and the poweramplifier.
 6. The RFID system of claim 1 further comprising a continuouswave source connected to the quadrature modulator.
 7. The RFID system ofclaim 1 wherein the envelope amplifier further comprises: a switch modepower supply; and a linear regulator connected to the switch mode powersupply.
 8. A method of operating a transmitter for a radio frequencyidentification (RFID) system, the method comprising: producing, using aprocessor, a cartesian in-phase signal and a cartesian quadrature-phasesignal; supplying the cartesian in-phase signal and the cartesianquadrature-phase signal to a quadrature modulator; supplying an envelopesignal from an envelope amplifier to a power amplifier; and driving thepower amplifier using the quadrature modulator to produce a transmitteroutput signal.
 9. The method of claim 8 further wherein driving thepower amplifier further comprises: using the quadrature modulator todrive a driver amplifier; and using the driver amplifier to drive thepower amplifier.
 10. The method of claim 8 further comprising supplyinga continuous wave signal to the quadrature modulator.
 11. The method ofclaim 8 wherein the transmitter output signal is produced using offsetphase reversal amplitude shift keying (OPR-ASK).
 12. Apparatus for radiofrequency identification (RFID) comprising: means for producing acartesian in-phase signal and a cartesian quadrature-phase signal; meansfor supplying an envelope signal to a power amplifier; and means formodulating the power amplifier using the cartesian in-phase signal andthe cartesian quadrature-phase signal the to produce a transmitteroutput signal.
 13. The apparatus of claim 12 further comprising meansfor supplying a continuous wave signal for use in modulating the poweramplifier.
 14. A radio frequency (RF) system comprising: a poweramplifier to produce a transmitter output signal; an envelope amplifierconnected to the power amplifier to supply an envelope signal to thepower amplifier; and a quadrature modulator connected to the poweramplifier to provide modulation for the transmitter output signal usinga cartesian input signal.
 15. The RF system of claim 14 wherein theenvelope amplifier further comprises a multi-phase buck converter. 16.The RF system of claim 14 further comprising a driver amplifierconnected to the power amplifier.
 17. The RF system of claim 16 furthercomprising a DC blocking capacitor connected between the driveramplifier and the power amplifier.
 18. The RF system of claim 14 furthercomprising a continuous wave source connected to the quadraturemodulator.
 19. The RF system of claim 14 wherein the envelope amplifierfurther comprises a switch mode power supply.
 20. The RF system of claim19 wherein the envelope amplifier further comprises a linear regulatorconnected to the switch mode power supply.